Method and apparatus for adapting a variable impedance network

ABSTRACT

The present disclosure may include, for example, a tunable capacitor having a decoder for generating a plurality of control signals, and an array of tunable switched capacitors comprising a plurality of fixed capacitors coupled to a plurality of switches. The plurality of switches can be controlled by the plurality of control signals to manage a tunable range of reactance of the array of tunable switched capacitors. Additionally, the array of tunable switched capacitors is adapted to have non-uniform quality (Q) factors. Additional embodiments are disclosed.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. application Ser. No.16/841,266, filed Apr. 6, 2020, which is a divisional of U.S.application Ser. No. 16/295,657, filed Mar. 7, 2019 (now U.S. Pat. No.10,615,769), which is a divisional of U.S. application Ser. No.15/673,613, filed Aug. 10, 2017 (now U.S. Pat. No. 10,263,595), which isa continuation of U.S. application Ser. No. 15/367,753, filed Dec. 2,2016 (now U.S. Pat. No. 9,742,375), which is a continuation of U.S.application Ser. No. 14/332,458, filed Jul. 16, 2014 (now U.S. Pat. No.9,548,716), which is a continuation of U.S. application Ser. No.12/729,221, filed Mar. 22, 2010 (now U.S. Pat. No. 8,803,631), which areincorporated herein by reference in their entirety.

FIELD OF THE DISCLOSURE

The present disclosure relates to variable impedance networks and morespecifically to a method and apparatus for adapting a variable impedancenetwork.

BACKGROUND

Existing multi-frequency wireless devices (e.g., radios) use an antennastructure that attempts to radiate at optimum efficiency over the entirefrequency range of operation, but can really only do so over a subset ofthe frequencies. Due to size constraints, and aesthetic design reasons,an antenna designer may be forced to compromise the performance in someof the frequency bands. An example of such a wireless device could be amobile telephone that operates over a range of different frequencies,such as 800 MHz to 2200 MHz. The antenna will not radiate efficiently atall frequencies due to the nature of the design, and the power transferbetween the antenna, the power amplifier, and the receiver in the radiocan vary a considerable amount.

Additionally, an antenna's performance can be impacted by its operatingenvironment. For example, multiple use cases exist for radio handsets,which include such conditions as the placement of the handset's antennanext to a user's head, in the user's pocket, the covering of an antennawith a hand, a pull-out antenna in the up position or down position, aflip phone with the lid open versus closed, hands-free operation with aBluetooth headset or speakerphone feature, or other operationalpossibilities, all of which can affect the wireless device antenna'sradiated efficiency.

Many existing radios use a simple circuit composed of fixed valuecomponents that are aimed at improving the power transfer from poweramplifier to antenna, or from the antenna to the receiver, but since thecomponents used are fixed in value there is typically a compromise whenattempting to cover multiple frequency bands and multiple use cases.

Prior art systems have attempted to solve this problem by employing avariety of tunable elements in the radio frequency path, thus attemptingto compensate for changing antenna performance. Typically, prior artsystem arrange these adjustable elements into single device substratesor semiconductor die, both to re-use control and bias circuitry betweenseveral tunable capacitors on the same die and to reduce the number ofinput/output connection pads necessary to connect the devices toexternal circuitry.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 depicts an illustrative embodiment of a communication device;

FIG. 2 depicts an illustrative embodiment of a portion of a transceiverof the communication device of FIG. 1;

FIGS. 3-4 depict illustrative embodiments of a tunable matching networkof the transceiver of FIG. 2;

FIGS. 5-6 depict illustrative embodiments of a tunable reactive elementof the tunable matching network;

FIG. 7 depicts an illustrative embodiment of a switched variablecapacitor array controlled by a switch decoder;

FIG. 8 depicts an illustrative embodiment of a simplified diagram of theswitched variable capacitor array of FIG. 7;

FIGS. 9-11 depict illustrative embodiments for improving a switchedvariable capacitor array;

FIG. 12 depicts an illustrative embodiment for integrating any of theimproved switched variable capacitor array of FIGS. 9-11 in a singleIntegrated Circuit (IC) or semiconductor die device;

FIGS. 13-17 depict illustrative embodiments of topologies applied to theIC device of FIG. 12;

FIGS. 18-20 depict illustrative embodiments of low band and high bandmatching networks;

FIG. 21 shows TABLE 1 which depicts an illustrative embodiment of atable with digital logic for controlling of a binary-switched capacitorarray; and

FIG. 22 shows TABLE 2 which depicts an illustrative embodiment of atable with digital logic for controlling a non-binary-switched capacitorarray.

DETAILED DESCRIPTION

An embodiment of the present disclosure can entail a tunable capacitorhaving a decoder for generating a plurality of control signals, and anarray of tunable switched capacitors comprising a plurality of fixedcapacitors coupled to a plurality of switches. The plurality of switchescan be controlled by the plurality of control signals to manage atunable range of reactance of the array of tunable switched capacitors.Additionally, the array of tunable switched capacitors is adapted tohave non-uniform quality (Q) factors.

An embodiment of the present disclosure can entail a variable matchingnetwork having a plurality of tunable capacitors contained on a singlesubstrate. The single substrate can have a plurality of ports coupled toportions of the plurality of tunable capacitors. Additionally, thevariable matching network can be configurable in a plurality oftopologies by way of the plurality of ports.

An embodiment of the present disclosure can entail a device having firstand second variable capacitors within a single substrate with each endof said first and second variable capacitors having an accessibleexternal port, wherein the first and second variable capacitors areadapted to be configured as a Tee, Pi or L circuit topology.

An embodiment of the present disclosure can entail a method for causingan array of tunable switched capacitors to adapt to a variable loadimpedance, wherein the array of tunable switched capacitors isconfigured to have non-uniform Q factors.

An embodiment of the present disclosure can entail a tunable capacitoron a die having a decoder for generating a plurality of control signals,and an array of tunable switched capacitors including a plurality offixed capacitors coupled to a plurality of switches. The plurality ofswitches can be controlled by the plurality of control signals to managea tunable range of reactance of the array of tunable switchedcapacitors. One of the plurality of fixed capacitors can be optionallycoupled to the array to reduce an aggregate parasitic capacitance of thearray of tunable switched capacitors. The array of tunable switchedcapacitors can be adapted to have uniform quality (Q) factors.

FIG. 1 depicts an exemplary embodiment of a communication device 100.The communication device 100 can comprise a wireless transceiver 102(herein having independent transmit and receiver sections, a userinterface (UI) 104, a power supply 114, and a controller 106 formanaging operations thereof. The wireless transceiver 102 can utilizeshort-range or long-range wireless access technologies such asBluetooth, WiFi, Digital Enhanced Cordless Telecommunications (DECT), orcellular communication technologies, just to mention a few. Cellulartechnologies can include, for example, CDMA-1X, WCDMA, UMTS/HSDPA,GSM/GPRS, TDMA/EDGE, EV/DO, WiMAX, and next generation cellular wirelesscommunication technologies as they arise.

The UI 104 can include a depressible or touch-sensitive keypad 108 witha navigation mechanism such as a roller ball, joystick, mouse, ornavigation disk for manipulating operations of the communication device100. The keypad 108 can be an integral part of a housing assembly of thecommunication device 100 or an independent device operably coupledthereto by a tethered wireline interface (such as a flex cable) or awireless interface supporting for example Bluetooth. The keypad 108 canrepresent a numeric dialing keypad commonly used by phones, and/or aQwerty keypad with alphanumeric keys. The UI 104 can further include adisplay 110 such as monochrome or color LCD (Liquid Crystal Display),OLED (Organic Light Emitting Diode) or other suitable display technologyfor conveying images to an end user of the communication device 100. Inan embodiment where the display 110 is a touch-sensitive display, aportion or all of the keypad 108 can be presented by way of the display.

The power supply 114 can utilize common power management technologies(such as replaceable batteries, supply regulation technologies, andcharging system technologies) for supplying energy to the components ofthe communication device 100 to facilitate portable applications. Thecontroller 106 can utilize computing technologies such as amicroprocessor and/or digital signal processor (DSP) with associatedstorage memory such a Flash, ROM, RAM, SRAM, DRAM or other liketechnologies.

FIG. 2 depicts an illustrative embodiment of a portion of the wirelesstransceiver 102 of the communication device 100 of FIG. 1. In GSMapplications, the transmit and receive portions of the transceiver 102can include common amplifiers 201, 203 coupled to a tunable matchingnetwork 202 and an impedance load 206 by way of a switch 204. The load206 in the present illustration can an antenna as shown in FIG. 1(herein antenna 206). A transmit signal in the form of a radio frequency(RF) signal (TX) can be directed to the amplifier 201 which amplifiesthe signal and directs the amplified signal to the antenna 206 by way ofthe tunable matching network 202 when switch 204 is enabled for atransmission session. The receive portion of the transceiver 102 canutilize a pre-amplifier 203 which amplifies signals received from theantenna 206 by way of the tunable matching network 202 when switch 204is enabled for a receive session. Other configurations of FIG. 2 arepossible for other types of cellular access technologies such as CDMA.These undisclosed configurations are contemplated by the presentdisclosure.

FIGS. 3-4 depict illustrative embodiments of the tunable matchingnetwork 202 of the transceiver 102 of FIG. 2. In one embodiment, thetunable matching network 202 can comprise a control circuit 302 and atunable reactive element 310. The control circuit 302 can comprise aDC-to-DC converter 304, one or more digital to analog converters (DACs)306 and one or more corresponding buffers 308 to amplify the voltagegenerated by each DAC. The amplified signal can be fed to one or moretunable reactive components 504, 506 and 508 such as shown in FIG. 5,which depicts a possible circuit configuration for the tunable reactiveelement 310. In this illustration, the tunable reactive element 310includes three tunable capacitors 504-508 and an inductor 502 with afixed inductance. Other circuit configurations are possible, and therebycontemplated by the present disclosure.

The tunable capacitors 504-508 can each utilize technology that enablestunability of the capacitance of said component. One embodiment of thetunable capacitors 504-508 can utilize voltage or current tunabledielectric materials such as a composition of barium strontium titanate(BST). An illustration of a BST composition is the Parascan® TunableCapacitor. In another embodiment, the tunable reactive element 310 canutilize semiconductor varactors. Other present or next generationmethods or material compositions that can support a means for a voltageor current tunable reactive element are contemplated by the presentdisclosure.

The DC-to-DC converter 304 can receive a power signal such as 3 Voltsfrom the power supply 114 of the communication device 100 in FIG. 1. TheDC-to-DC converter 304 can use common technology to amplify this powersignal to a higher range (e.g., 30 Volts) such as shown. The controller106 can supply digital signals to each of the DACs 306 by way of acontrol bus of “n” or more wires to individually control the capacitanceof tunable capacitors 504-508, thereby varying the collective reactanceof the tunable matching network 202. The control bus can be implementedwith a two-wire common serial communications technology such as a SerialPeripheral Interface (SPI) bus. With an SPI bus, the controller 106 cansubmit serialized digital signals to configure each DAC in FIG. 3 or theswitches of the tunable reactive element 404 of FIG. 4. The controlcircuit 302 of FIG. 3 can utilize common digital logic to implement theSPI bus and to direct digital signals supplied by the controller 106 tothe DACs.

In another embodiment, the tunable matching network 202 can comprise acontrol circuit 402 in the form of a decoder and a tunable reactiveelement 404 comprising switchable reactive elements such as shown inFIG. 6. In this embodiment, the controller 106 can supply the controlcircuit 402 signals via the SPI bus which can be decoded with commonBoolean or state machine logic to individually enable or disable theswitching elements 602. The switching elements 602 can be implementedwith semiconductor switches or micro-machined switches such as utilizedin micro-electromechanical systems (MEMS). By independently enabling anddisabling the reactive elements 604 (capacitor or inductor) of FIG. 6with the switching elements 602, the collective reactance of the tunablereactive element 404 can be varied.

The tunability of the tunable matching networks 202, 204 provides thecontroller 106 a means to optimize performance parameters of thetransceiver 102 such as, for example, but not limited to, transmitterpower, transmitter efficiency, receiver sensitivity, power consumptionof the communication device, a specific absorption rate (SAR) of energyby a human body, frequency band performance parameters, and so on. Toachieve one or more desirable performance characteristics which can bedefined, the communication device 100 utilizes a tuning state selectionmethod as depicted in FIG. 7.

FIG. 7 depicts an illustrative embodiment of a switched variablecapacitor array controlled by a switch decoder. The switched variablecapacitor array can collectively represent one variable capacitor havinga tuning range of operation which depends on the switching logic used toadd and remove the fixed capacitors (C₁-C₅) shown in the top portion.Each of the fixed capacitors C₁-C₅ can be removed or applied to thearray by a semiconductor (or MEMS) switch (S₁) controlled by a switchdecoder that utilizes common Boolean and/or state machine logic toenable or disable the switches according to a predetermined tuningrange. Each switch (S₁) has parasitic on-resistance and off-capacitancethat can be controlled by an integrated circuit designer by the channelsize of the switch and by the number of parallel and/or seriessemiconductor (or MEMS) switches used. The structure and size of eachswitch (S₁) determines the magnitude of these parasitics and the affectit has on the operating characteristics of the switched variablecapacitor array as will be discussed shortly. For ease of discussion,references may be made to FIG. 8 which depicts an illustrativeembodiment of a simplified diagram of the switched variable capacitorarray of FIG. 7 without the switch decoder.

FIGS. 9-11 depict illustrative embodiments for improving the switchedvariable capacitor array of FIG. 8. Note that although FIGS. 9-11 do notshow the parasitic capacitance depicted in FIGS. 7-8, such capacitanceis inherently present in switches S₁-S₅, but not shown to simplify theillustration in these figures.

One embodiment of the present disclosure can entail changing theconfiguration and connections of the binary array of switchablecapacitors such that some portions of the array, typically the largestcapacitor in the array (C₅), can be left unconnected from the substrateif the larger value is not required by a specific tunable matchingnetwork for a specific application. For instance a typical switchablecapacitor binary array can have the following values in the array: ¼ pF,½ pF, 1 pF, 2 pF and 4 pF (shown in FIGS. 9-11). This array cantheoretically be tunable between 0 pF and 7¾ pF by either connecting ordisconnecting none, some or all of the capacitors in the array to thetunable matching network. However, there are practical limitations tothe minimum capacitance which can be achieved in the actual fabricationof such a binary array.

Specifically, the switches themselves can have a certain amount ofparasitic off-capacitance as shown in FIG. 8 which is present when theswitch is in the “open” configuration, thus this parasitic capacitancelimits the minimum capacitance value of the array to a value larger thanzero. It is common for each individual capacitor-switch combination(e.g., C₁-S₁) shown in FIGS. 9-11 to have parasitic resistance andcapacitance (C_(p), R_(p)) shown in the switch combination S_(n) of FIG.7. As the switches are made larger, which is necessary as the switchedcapacitance gets larger (to maintain circuit quality factor Q and reducelosses), this parasitic capacitance grows proportionally. Thus, arraysthat have large values of capacitance and are able to adjust to highertotal capacitance values are inherently limited in how low they can alsoadjust.

For illustration purposes, suppose the parasitic capacitance of eachcapacitor-switch combination is ¼ the value of the fixed capacitor ofthe capacitor-switch combination. One can gather that the parasiticcapacitance of the first capacitor-switch combination would be 1/16 pF,the second capacitor-switch combination ⅛ pF, the third capacitor-switchcombination ¼ pF, the fourth capacitor-switch combination ½ pF, and thefifth capacitor-switch combination 1 pF. When all switches are in theoff state, the aggregate off capacitance is 1.9375 pF which can beapproximated to 2 pF. The maximum capacitance of the array of fivecapacitors is 7.75 pF (¼+½+1+2+4) which can be approximated to 8 pF.Taking into consideration parasitic capacitance in the off state, andthe total capacitance of all fixed capacitors when engaged, one cansurmise that the binary array of FIG. 9 as a 4:1 tuning ratio. Theparasitic capacitance of the largest corresponding switch (S₅) can add 1pF which limits the minimum value of the array to 2 pF, therebyincreasing the minimum capacitance value available to the circuit.

One aspect of a switched capacitor is that the circuit quality or Q ofthe switched capacitor includes the resistive losses in the MEMS orsemiconductor switch which is in series with the capacitor. The Q of theresultant switched capacitor is the ratio of the reactance of thecapacitor to the resistance of the combined switch and capacitor whichwe will define as R_(total).R _(total) =R _(sw) +R _(cap)

Where R_(sw) is the resistance of the switch connected to a particularcapacitor and R_(cap) is the effective series resistance of theparticular capacitor in the array. The reactance of the capacitor, X_(C)is:

${{Xc} = \frac{1}{\omega\; C}},$

where ω=2πf

where f is the frequency of operation. Therefore, the Q of the switchedcapacitor is:

$Q = {\frac{X_{c}}{R_{total}} = {\frac{1\text{/}\omega\; C}{R_{total}} = {\frac{1}{\omega\; R_{total}C} = {\frac{1}{2\pi\;{fR}_{total}C} = \frac{1}{2\pi\;{f\left( {R_{sw} + R_{cap}} \right)}C}}}}}$

A similar analysis can be done wherein the switched reactances areinductors instead of capacitors. In that case, the reactance of theinductor, X_(L) is:X _(L) =ωL,

where ω=2πf

where f is the frequency of operation. Therefore, the Q of the switchedinductor is:

$Q = {\frac{X_{L}}{R_{total}} = {\frac{\omega\; L}{R_{total}} = {\frac{2\pi\;{fL}}{R_{total}} = \frac{2\pi\;{fL}}{\left( {R_{sw} + R_{cap}} \right)}}}}$

Correspondingly, in order to maintain Q at a given frequency ofoperation, the resistance of the switch in the array must be inverselyproportional to the size of the inductor in the array. Therefore, if aninductor in the array is one half the size of the next inductor in thearray, the resistance of that switch must also be one half theresistance of the switch for the preceding inductor. As such, thesmallest inductor in the array must have the largest switch (in order toreduce resistance and hence maintain Q) which is the oppositerelationship of the switchable capacitor array. However, although therelationship is reversed by virtue of the nature of the reactance of theswitched component, it is understood that all of the characteristics andattributes being described herein can be applied to either capacitors orinductors with this simple modification. What follows are illustrativeembodiments of tunable switched capacitor arrays. However, a tunableswitched inductor array is also contemplated by the present disclosure.

Typically, for size and cost reasons the fixed capacitors in the arraywill be situated within the semiconductor die and will be built usingtypical semiconductor capacitor fabrication methods such asmetal-insulator-metal (MIM) or other known technologies. Thesesemiconductor capacitor technologies are typically limited in the Qvalues they are capable of achieving, and as seen in the expressionsabove, the Q of these fixed capacitors has a significant impact on thetotal Q of the tunable capacitor array, and as such force the designerto size the switches large enough to compensate for the limited Q of thefixed capacitors. This limitation has the most impact on the die size ofthe switch connected to the largest value capacitor in the array.

Replacing this largest value capacitor with an external fixed capacitorwould allow the designer to utilize a capacitor with a Q perhaps twicethe Q available from semiconductor capacitors, and by doing so wouldallow the designer to reduce the size of the largest switch in the arraysignificantly—see FIG. 11. For example, if the designer needed tomaintain a Q factor for the switched capacitor of 60, by using anexternal capacitor with a Q of 250 instead of an internal semiconductorcapacitor with a Q of 150 the designer would be able to increase theresistance of the switch on that capacitor by over 25%, which wouldallow the die area of that particular switch to be about 20% smaller.Since the switch for the largest capacitor in the binary array usuallyaccounts for about ½ of the total switch die area of the array, thisarea savings would amount to approximately 10% of the total array switcharea.

As was stated previously, in a typical switched capacitor binary array,the individual capacitor values are selected such that each capacitor istwice the capacitance of the previous capacitor in the array, with thesmallest capacitor being the size required for the minimum resolution ofthe tunable capacitor. One can see that as the value of C increases, inorder to maintain a particular Q value the resistance of the switchconnected to the capacitor has to be decreased inversely proportionally.From further examination of the expression for Q, it can also be seenthat the Q of the switched capacitor is inversely proportional to theoperating frequency, such that the minimum Q value is apparent at thehighest operating frequency of the capacitor. In other words, the switchmust be designed to have a resistance small enough to maintain thetarget Q value at the highest frequency of operation.

This characteristic has an aspect that can be taken advantage of when atunable capacitor such as shown in FIGS. 7-11 is utilized in a matchingnetwork of a cellular handset that operates over a wide frequency rangesuch as, for example, 850 MHz, 900 MHz, 1800 MHz and 1900 MHz frequencybands. For convenience we will refer to the 850 MHz and 900 MHz bands asthe “low bands” as they are both below 1 GHz and the 1800 MHz and 1900MHz bands as the “high bands” as they are both just below 2 GHz.Considering the above expression for the Q of a switched capacitor, itis evident that in order to maintain the same value of Q, the resistanceof a switch would need to be approximately half as large for a capacitorof a given value operating in the high bands as it would be for the samecapacitor operating in the low bands. To accomplish this on a MEMS orsemiconductor switch die would correspondingly require twice the diearea to create the switch for the capacitor to operate in the high bandas it would to operate in the low band.

It is another characteristic typical of tunable matching networks thatlarger value capacitors are usually only required when operating in thelow band, while smaller capacitors are typically utilized when operatingin the high band. This is primarily due to the fact that the reactanceis an inverse function of the operating frequency as seen above, and asfrequency increases, large value capacitors become very low inreactance, and as such have little effect on tuning or matching at highband frequencies when they are placed in a circuit in a seriesconfiguration, and would have too large an effect on tuning or matchingwhen placed in a circuit in a shunt configuration.

This characteristic can be exploited in one embodiment of the presentdisclosure whereby the largest capacitor in a switchable capacitorbinary array could be switched by a semiconductor or MEMS switch thathad a larger resistance than that which would be required to maintainthe target Q value in the high band, but instead sized to only maintainthe target Q value in the low band—see FIG. 9. Doing so wouldsignificantly reduce the die area of the switch associated with thelarge value capacitor, and correspondingly reduce the cost of the die.Given the approximate 2 to 1 relationship of the high band and low bandfrequencies, allowing the switch to be one half the size (twice theresistance) for the largest value capacitor would result in anapproximate 25% savings in the die area required for the totalswitchable capacitor binary array, since the largest value capacitor inthe binary array is approximately one half of the total capacitance inthe array.

An embodiment of the present disclosure can be applied to tuning asingle reactive circuit element such as an antenna or other resonantstructure. In such cases, the tunable capacitor can operate in thecircuit along with another reactive element such as an inductor, and inpractice the circuit will be tuned in order to be resonant at thefrequency of operation. By virtue of the tunable capacitance, such acircuit can be tuned over a range of frequencies.

As an example, the analysis below considers a resonant circuitcontaining an inductor of value L and capacitor of value C, for whichthe equation below describes the formula for the resonant frequency.ω_(res)=2πf _(res)=1/√{square root over (LC)}

As the expression shows, the resonant frequency is inverselyproportional to the square root of the value of the capacitor, and whenthe value of the tunable capacitor is adjusted to achieve resonance atthe desired frequency of operation, the value of the tunable capacitorwill then be inversely proportional to the square of the desiredfrequency of operation. Correspondingly, at higher frequencies ofoperation, the capacitor will be tuned to lower values. Since the Q ofthe switched capacitor is an inverse function of C as was seen above,one can deduce that for each higher individual capacitor in a binaryweighted array, the resistance will not need to be reduced by the samefactor as the capacitance was increased. Therefore, while in order tomaintain uniform Q of a tunable capacitor the switch resistance isusually cut in half (and switch die area doubled) for each doubling incapacitor size, for applications in which it is known that operatingfrequency is inversely proportional to the square root of thecapacitance value the switch resistance can be made larger for largercapacitors in the array, thereby saving additional die area and cost.

As was shown above:

$Q = \frac{1}{2\pi\;{fR}_{total}C}$

Substituting for f to obtain an expression for Q_(res) at f_(res):

$Q_{res} = {\frac{1}{2\pi\; f_{res}{RC}} = {\frac{1}{2\pi\;{RC}\text{/}2\pi\sqrt{LC}} = \frac{\sqrt{L}}{R\sqrt{C}}}}$

Solving for R at f_(res):

$R = \frac{\sqrt{L}}{Q\sqrt{C}}$

And, if Q is maintained as f_(res) is tuned to the frequency ofoperation:

$R \propto \frac{1}{\sqrt{C}}$

Therefore, in order to maintain circuit Q in such an application, thetotal resistance for a capacitor in the array which is two times thesize of another capacitor in the array need only have the resistancereduced by a factor of 1.414 (or to 71% instead of 50%).Correspondingly, in an exemplary application, for the largest capacitorin a binary array, the size of the switch may be only 1.414 times thesize of the switch for the next largest capacitor in the array insteadof normally being twice the size (in an application where Q ismaintained uniformly across all capacitor values at a given frequency).Utilizing an array with non-uniform Q can result in a 29% reduction inthe size of this switch, and by applying this principal to the entirebinary array, a reduction of approximately the same percentage of thetotal switch area can be accomplished.

While this analysis is representative for a tuned circuit that includesa simple resonant inductor-capacitor circuit, for other circuits theresonance may not follow the exact f_(res)=1/2π√{square root over (LC)}relationship. However, in general, the circuit will typically requirelarger capacitance values at lower frequencies of operation, and oncethe exact relationship is determined, the appropriate scaling factor forthe switch resistances required to achieve the required Q values can bedetermined and applied to the switched capacitor array.

A further extension of the above embodiment is to build the switchedcapacitor array in a non-binary fashion. That is, to not have eachcapacitor (except the smallest capacitors) in the array be one-half thecapacitance of another capacitor in the array, but to rather have anarray which splits the tunable capacitor into differentnon-binary-weighted capacitors. For example, an array of ¼ pF, ½ pF, 1pF, 1 pF, 1 pF, can have the same tuning range as the binary weightedswitched capacitor array ¼ pF, ½ pF, 1 pF, 2 pF, and can also have thesame minimum resolution of ¼ pF (see FIG. 21, TABLE 1 and FIG. 22, TABLE2). However, by further exploiting the characteristic explained aboveand allowing the Q of each individual capacitor and switch combinationto vary, further reduction in total switch die size can be accomplished,and correspondingly, further reduction in die cost can be achieved.Since in some applications as described above, as the operatingfrequency is increased the tunable capacitor will typically be set tolower values of capacitance, each switch connected to the array of 1 pFcapacitors (in this example) can be made of different size such that thetotal Q is maintained in the application. By allowing each 1 pFcapacitor to individually be paired with a smaller switch, andactivating them in a particular order, the Q of the total capacitancebeing switched into the tunable circuit can be maintained up to thefrequency of operation.

In another embodiment this disadvantage can be overcome by allowing thelargest value capacitor in the array (in the present illustration, the 4pF capacitor) to be connected separately from the array by way of a portin the substrate as shown in FIGS. 10-11. In this way, if the design ofthe tunable capacitor requires a smaller total capacitance, the largestvalue in the array can be left unattached from the array, therebyallowing the tunable capacitor array to achieve smaller values ofcapacitance than if the largest capacitor was included in the circuit.In this embodiment, a circuit designer can choose between connecting thefour smallest capacitors (C₁-C₄) which could give a tuning range of 1 pF(the approximate off capacitance of S1-S4, i.e., 1/16+⅛+¼+½) to 4 pF(4:1 tuning), or connecting all five capacitors which can give a tuningrange of 2 pF to 8 pF (4:1 tuning) as described earlier. Both choicesare available on the same die. If the largest capacitor in the arraywere not separated from the array as shown in FIGS. 10 and 11, adesigner would be limited by a minimum value of 2 pF when the array isin the “off” state—assuming the switch size of the largest capacitorfollowed a binary array.

One embodiment of the present disclosure can entail a single die such asshown in FIG. 12 that includes multiple tunable capacitors that can beplaced onto circuit substrates which allow the tunable capacitors on thedie to be connected into several different tunable matching networktopologies which can be completely defined by the connecting metalpatterns on the circuit substrates. It should be noted that the tuningrange of each of the tunable capacitors can be controlled by a serialbus which feeds an independent decoder integrated with each tunablecapacitor. These substrates can be multi-layer printed circuit boards,or multi-layer ceramic substrates and allow the die to connect to otherelectronic components such as fixed value capacitors and inductors whichtogether embody the tunable matching network. Such substrates aretypically very inexpensive, and as such multiple versions, embodyingmultiple schematic topologies can be kept in stock by a tunable matchingnetwork manufacturer, thus allowing the flexibility of quicklyconverting many different tunable matching network designs to productiononce the final design is determined.

One embodiment of the present disclosure can entail the inclusion offixed value capacitors on the substrates which would connect in parallelwith the tunable capacitors on the die to provide additional capacitorvalue range for the tunable matching network design. For instance, thetunable capacitor on the die may have the range of values from 1picofarads to 5 picofarads, but by adding a fixed 2 picofarad capacitorin parallel, the effective range of the capacitor could be extended to 3picofarads to 7 picofarads. This would reduce the tuning ratio from 5:1to 7:3, but by extending the value provides flexibility which may berequired to accomplish the matching impedances required for a specificapplication without changing the die which embodies the tunablecapacitors.

FIG. 12 depicts a block diagram of a single die that could contain 3tunable capacitors connected in a manner that would allow the die to beconnected in either a “Tee” configuration as shown in FIGS. 13 and 16,or a “Pi” configuration as shown in FIGS. 14 and 17, or an “L”configuration as shown in FIG. 15. It should be noted that tunablecapacitor 1202 of FIG. 12 could be connected in parallel or in serieswith either of the other tunable capacitors on the die to create atunable capacitor with either a larger total value or smaller(respectively) while still maintaining tuning ratio of the combinedcapacitance. Also note that a myriad of circuit topologies could becreated with the three tunable capacitors on the die, and the modulesubstrate is all that needs to change for this single die to support allthese possible topologies. For example, an external fixed capacitor canbe connected in parallel to one of the tunable capacitors on the die ofFIG. 16 to increase the effective value of the tunable capacitor inparallel with said fixed capacitor.

Modern wireless systems need to support multiple modes and bands ofoperation. There can be multiple ways to architect a radio system tosupport such systems. In certain cases, antenna or other tunablematching networks may be deployed in separate circuit paths. Onespecific example would be to separate the circuit paths by frequencyrange, such as low band and high band ranges. Another example would beto separate the circuit paths transmit and receive. The illustrations inFIGS. 18-20 are specific to a high band—low band separation—additionalembodiments are contemplated.

FIG. 18 depicts low band and high band independent matching networks ona single die. FIG. 19 depicts an embodiment in which signal switches areintegrated into the tunable matching network die to allow the matchingnetwork to be divided into two separate paths. One or both paths caninclude tunable matching networks. One of the switches on one side ofthe matching networks of FIG. 19 could be eliminated while the switch onthe other port is retained, which would allow such a tuner to be placedin front of a single input switchplexer (radio) but connected to a dualfeed antenna, or two separate antennas. Alternatively, the switch on oneside of the matching networks of FIG. 19 could be eliminated while theswitch on the other port is retained, which would allow such a tuner tobe placed in front of a single input antenna but connected to a dualinput switchplexer (radio).

FIG. 20 depicts an embodiment that introduces additional flexibility tothe integrated circuit die which embodies this circuit by allowing it tobe used in both single path architectures and dual path architectures.It should be considered that a switch in FIG. 20 could be eliminated byfixing it in one position, allowing the tunable matching network tointerface to a handset front end switchplexer that was configured tohave two inputs rather than just a single antenna input. It should alsobe considered that a switch in FIG. 20 could be eliminated by fixing itin one position, allowing the tunable matching network to interface to ahandset antenna that was configured to have two inputs rather than justa single antenna input.

In sum, one embodiment of the present disclosure can entail theinclusion of additional switches on the semiconductor or MEMS die of amatching network with the function of separating the signal paths withinthe tunable matching network. Such an approach can allow the designer toprovide for separate tunable matching networks for either the low bandor high band by switching the signal path at the input and outputs oftwo separate tunable matching networks, diverting the signal to one orthe other depending upon the frequency of operation, or any otherattribute by which the two paths could be distinguished. In someinstances, one of the paths could consist of a simple “through”connecting the input to the output, or a fixed matching network therebyproviding a tunable matching network in one band and a fixed matchingnetwork in the other band.

Referring to FIG. 17, one embodiment of the present disclosure canentail the inclusion of additional switches on the semiconductor or MEMSdie of a matching network with the function of bypassing certainsections of the tunable impedance match. Certain tunable matchingnetworks may include sections that could include for instance a seriestunable capacitor and series inductor which may provide significantimpedance tuning in the low band, but introduce a high impedance in thehigh band and as such introduce significant insertion loss in the highband. Bypassing, or shunting around these two parts could allow thetunable matching network to have much lower loss in the high band byeffectively removing those components when they are not required. Othersuch circuit elements or sections of circuitry could be bypassed forsimilar reasons in either the low band or high band, or in some caseswhen matching particular impedances which do not require the specificelements being bypassed. This is an example of such an application butis not intended to limit the applications of such a shunt/bypass switchfor use in tunable matching networks.

Generally speaking, to maximize the performance of a tunable matchingnetwork, and also to reduce the total cost, a typical tunable matchingnetwork should be designed uniquely for a specific application, and todo so would require a unique tunable capacitor die to be designed foreach application. Experience shows that in the case where theapplication is a tunable matching network for a cellular handsetantenna, this would require significant delays in the production of sucha handset, since the tunable capacitor die would not be defined untilafter the handset antenna design was completed which is typically towardthe end of the handset design cycle. Correspondingly, it would be verydifficult for a handset designer to meet what are usually veryaggressive schedule targets and use a tunable matching network thatrequired a unique tunable capacitor die due to the issues describedabove. The various embodiments described in the present disclosure andothers contemplated by the scope of the claims below clearly overcomethe disadvantages of present systems.

The illustrations of embodiments described herein are intended toprovide a general understanding of the structure of various embodiments,and they are not intended to serve as a complete description of all theelements and features of apparatus and systems that might make use ofthe structures described herein. Many other embodiments will be apparentto those of skill in the art upon reviewing the above description. Otherembodiments may be utilized and derived therefrom, such that structuraland logical substitutions and changes may be made without departing fromthe scope of this disclosure. Figures are also merely representationaland may not be drawn to scale. Certain proportions thereof may beexaggerated, while others may be minimized. Accordingly, thespecification and drawings are to be regarded in an illustrative ratherthan a restrictive sense.

Such embodiments of the inventive subject matter may be referred toherein, individually and/or collectively, by the term “invention” merelyfor convenience and without intending to voluntarily limit the scope ofthis application to any single invention or inventive concept if morethan one is in fact disclosed. Thus, although specific embodiments havebeen illustrated and described herein, it should be appreciated that anyarrangement calculated to achieve the same purpose may be substitutedfor the specific embodiments shown. This disclosure is intended to coverany and all adaptations or variations of various embodiments.Combinations of the above embodiments, and other embodiments notspecifically described herein, will be apparent to those of skill in theart upon reviewing the above description.

What is claimed is:
 1. A device comprising: a tunable matching networkon a single die, the tunable matching network including a plurality ofsignal switches, wherein the plurality of signal switches areconfigurable to provide a first matching network and a separate secondmatching network having respectively a low-frequency signal path and ahigh-frequency signal path, wherein the single die has a plurality ofports for coupling to an input and an output of the first matchingnetwork and to an input and an output of the second matching network;wherein one or both of the low-frequency signal path and thehigh-frequency signal path comprises a tunable network; and wherein aninput switch of the plurality of signal switches is directly coupled toan input port of the plurality of ports and an output switch of theplurality of signal switches is directly coupled to an output port ofthe plurality of ports.
 2. The device of claim 1, wherein at least aportion of the plurality of signal switches are integrated into the die.3. The device of claim 1, wherein one or both of the low-frequencysignal path and the high-frequency signal path comprises a tunablenetwork; wherein a first input port of the plurality of ports and aseparate second input port of the plurality of ports are respectivelycoupled to the input of the first matching network and the input of thesecond matching network, and wherein the output switch of the pluralityof signal switches is configured to connect the output port alternatelyto the output of the first matching network or the output of the secondmatching network.
 4. The device of claim 3, wherein the first input portand the second input port are connected to a dual-mode antenna.
 5. Thedevice of claim 1, wherein the device comprises a radio system, andwherein the tunable matching network is configured to connect to one ofa single-input radio or a dual-input radio.
 6. The device of claim 1,wherein the device comprises a handset, and wherein the tunable matchingnetwork is configured to interface to an antenna of the handset.
 7. Thedevice of claim 1, wherein a first output port of the plurality of portsand a separate second output port of the plurality of ports arerespectively coupled to the output of the first matching network and theoutput of the second matching network, and wherein the input switch ofthe plurality of signal switches is configured to connect the input portalternately to the input of the first matching network or the input ofthe second matching network.
 8. The device of claim 7, wherein the inputswitch is configured to divert a signal to one of the low-frequencysignal path or the high-frequency signal path, in accordance with afrequency of the signal.
 9. The device of claim 1, wherein the device isconfigured to provide a tunable matching network in one of alow-frequency signal band and a high-frequency signal band and a fixedmatching network in the other of the low-frequency signal band and thehigh-frequency signal band.
 10. A device comprising: a tunable matchingnetwork on a single die, the tunable matching network including aplurality of signal switches, wherein the plurality of signal switchesare configurable to provide a first tunable matching network and aseparate second tunable matching network having respectively alow-frequency signal path and a high-frequency signal path, and todivert a signal to one of the low-frequency signal path or thehigh-frequency signal path, in accordance with a frequency of thesignal; and wherein an input switch of the plurality of signal switchesis directly coupled to an input port of the plurality of ports and anoutput switch of the plurality of signal switches is directly coupled toan output port of the plurality of ports.
 11. The device of claim 10,wherein at least a portion of the plurality of signal switches areintegrated into the die.
 12. The device of claim 10, wherein the singledie has a plurality of ports for coupling to an input and an output ofthe first tunable matching network and to an input and an output of thesecond tunable matching network.
 13. The device of claim 12, wherein afirst input port of the plurality of ports and a separate second inputport of the plurality of ports are respectively coupled to the input ofthe first tunable matching network and the input of the second tunablematching network, and wherein the output switch of the plurality ofsignal switches is configured to connect the output port alternately tothe output of the first tunable matching network or the output of thesecond tunable matching network.
 14. The device of claim 13, wherein thefirst input port and the second input port are connected to a dual-modeantenna.
 15. The device of claim 12, wherein a first output port of theplurality of ports and a separate second output port of the plurality ofports are respectively coupled to the output of the first tunablematching network and the output of the second tunable matching network,and wherein the input switch of the plurality of signal switches isconfigured to connect the input port alternately to the input of thefirst tunable matching network or the input of the second tunablematching network.
 16. A device comprising: a tunable matching network ona single die, the tunable matching network including a plurality ofsignal switches, wherein the plurality of signal switches areconfigurable to provide in the tunable matching network a first signalpath and a separate second signal path, the first signal path and thesecond signal path respectively comprising a low-frequency signal pathand a high-frequency signal path, wherein the single die has a pluralityof ports for coupling to an input and an output of the first signal pathand to an input and an output of the second signal path; wherein theplurality of signal switches are configured to divert a signal to one ofthe low-frequency signal path or the high-frequency signal path, inaccordance with a frequency of the signal; and wherein an input switchof the plurality of signal switches is directly coupled to an input portof the plurality of ports and an output switch of the plurality ofsignal switches is directly coupled to an output port of the pluralityof ports.
 17. The device of claim 16, wherein at least a portion of theplurality of signal switches are integrated into the die.
 18. The deviceof claim 16, wherein a first input port of the plurality of ports and aseparate second input port of the plurality of ports are respectivelycoupled to the input of the first signal path and the input of thesecond signal path, and wherein the output switch of the plurality ofsignal switches is configured to connect the output port alternately tothe output of the first signal path or the output of the second signalpath.
 19. The device of claim 18, wherein the first input port and thesecond input port are connected to a dual-mode antenna.
 20. The deviceof claim 16, wherein a first output port of the plurality of ports and aseparate second output port of the plurality of ports are respectivelycoupled to the output of the first signal path and the output of thesecond signal path, and wherein the input switch of the plurality ofsignal switches is configured to connect the input port alternately tothe input of the first signal path or the input of the second signalpath.